High speed communications system

ABSTRACT

Transmission of baseband and carrier-modulated vector codewords, using a plurality of encoders, each encoder configured to receive information bits and to generate a set of baseband-encoded symbols representing a vector codeword; one or more modulation circuits, each modulation circuit configured to operate on a corresponding set of baseband-encoded symbols, and using a respective unique carrier frequency, to generate a set of carrier-modulated encoded symbols; and, a summation circuit configured to generate a set of wire-specific outputs, each wire-specific output representing a sum of respective symbols of the carrier-modulated encoded symbols and at least one set of baseband-encoded symbols.

This application is a continuation of U.S. application Ser. No.15/194,497, filed Jun. 27, 2016, entitled “High Speed CommunicationsSystem”, which is a Non-Provisional Application claiming priority under35 USC § 119 to U.S. Provisional Application No. 62/189,953, filed Jul.8, 2015, entitled “High Speed Communications System”, and also claimingpriority under 35 USC § 119 to U.S. Provisional Application 62/185,403,filed Jun. 26, 2015, entitled “High Speed Communications System,” all ofwhich are hereby incorporated herein by reference in their entirety forall purposes.

REFERENCES

The following references are herein incorporated by reference in theirentirety for all purposes:

U.S. Patent Publication No. 2011/0268225 of U.S. patent application Ser.No. 12/784,414, filed May 20, 2010, naming Harm Cronie and AminShokrollahi, entitled “Orthogonal Differential Vector Signaling”,hereinafter identified as [Cronie I];

U.S. patent application Ser. No. 13/030,027, filed Feb. 17, 2011, namingHarm Cronie, Amin Shokrollahi and Armin Tajalli, entitled “Methods andSystems for Noise Resilient, Pin-Efficient and Low Power Communicationswith Sparse Signaling Codes”, hereinafter identified as [Cronie II];

U.S. patent application Ser. No. 14/158,452, filed Jan. 17, 2014, namingJohn Fox, Brian Holden, Peter Hunt, John D Keay, Amin Shokrollahi,Richard Simpson, Anant Singh, Andrew Kevin John Stewart, and GiuseppeSurace, entitled “Chip-to-Chip Communication with Reduced SSO Noise”,hereinafter identified as [Fox I];

U.S. patent application Ser. No. 13/842,740, filed Mar. 15, 2013, namingBrian Holden, Amin Shokrollahi and Anant Singh, entitled “Methods andSystems for Skew Tolerance in and Advanced Detectors for VectorSignaling Codes for Chip-to-Chip Communication”, hereinafter identifiedas [Holden I];

U.S. Provisional Patent Application No. 61/934,804, filed Feb. 2, 2014,naming Ali Hormati and Amin Shokrollahi, entitled “Methods for CodeEvaluation Using ISI Ratio”, hereinafter identified as [Hormati I];

U.S. Provisional Patent Application No. 62/026,860, filed Jul. 21, 2014,naming Ali Hormati and Amin Shokrollahi, entitled “Multidrop DataTransfer”, hereinafter identified as [Hormati II];

U.S. Provisional Patent Application No. 61/934,807, filed Feb. 2, 2014,naming Amin Shokrollahi, entitled “Vector Signaling Codes with Highpin-efficiency and their Application to Chip-to-Chip Communications andStorage”, hereinafter identified as [Shokrollahi I];

U.S. Provisional Patent Application No. 61/839,360, filed Jun. 23, 2013,naming Amin Shokrollahi, entitled “Vector Signaling Codes with ReducedReceiver Complexity”, hereinafter identified as [Shokrollahi II].

U.S. Provisional Patent Application No. 61/946,574, filed Feb. 28, 2014,naming Amin Shokrollahi, Brian Holden, and Richard Simpson, entitled“Clock Embedded Vector Signaling Codes”, hereinafter identified as[Shokrollahi III].

U.S. Provisional Patent Application No. 62/015,172, filed Jul. 10, 2014,naming Amin Shokrollahi and Roger Ulrich, entitled “Vector SignalingCodes with Increased Signal to Noise Characteristics”, hereinafteridentified as [Shokrollahi IV].

U.S. patent application Ser. No. 13/895,206, filed May 15, 2013, namingRoger Ulrich and Peter Hunt, entitled “Circuits for Efficient Detectionof Vector Signaling Codes for Chip-to-Chip Communications using Sums ofDifferences”, hereinafter identified as [Ulrich I].

“Controlled Intersymbol Interference Design Techniques of ConventionalInterconnection Systems for Data Rates beyond 20 Gbps”, WendemagegnehuT. Beyene and Amir Amirkhany, IEEE Transactions on Advanced Packaging,Vol. 31 No. 4, pg. 731-740, November 2008, hereinafter identified as[Beyene].

TECHNICAL FIELD

The present invention relates to communications in general and inparticular to the transmission of signals capable of conveyinginformation and detection of those signals in wired communication.

BACKGROUND

In communication systems, a goal is to transport information from onephysical location to another. It is typically desirable that thetransport of this information is reliable, is fast and consumes aminimal amount of resources. Methods of information transport arebroadly categorized into “baseband” methods that dedicate use of thephysical communications channel to one transport method, and “broadband”methods that partition the physical communications channel in thefrequency domain, creating two or more independent frequency channelsupon which a transport method may be applied.

Baseband methods may be further categorized by physical medium. Onecommon information transfer medium is the serial communications link,which may be based on a single wire circuit relative to ground or othercommon reference, multiple such circuits relative to ground or othercommon reference, or multiple such circuits used in relation to eachother. A common example of the latter uses differential signaling(“DS”). Differential signaling operates by sending a signal on one wireand the opposite of that signal on a matching wire. The signalinformation is represented by the difference between the wires, ratherthan their absolute values relative to ground or other fixed reference.

Parallel data transfer is also commonly used to provide increasedinterconnection bandwidth, with busses growing from 16 or fewer wires,to 32, 64, and more. As crosstalk and noise induced on the parallelsignal lines can produce receive errors, parity was added to improveerror detection, and signal anomalies were addressed through active bustermination methods. However, these wide data transfer widths inevitablyresulted in data skew, which became the limiting factor in increased busdata transfer throughput. Alternative approaches were developedutilizing narrower bus widths operating at much higher clock speeds,with significant effort placed on optimizing the transmission linecharacteristics of the interconnection medium, including use ofimpedance-controlled connectors and micro stripline wiring. Even so, theinevitable path imperfections required use of active equalization andinter-symbol interference (ISI) elimination techniques, including activepre-emphasis compensation for transmitters and Continuous Time LinearEqualization (CTLE) and Decision Feedback Equalization (DFE) forreceivers, all of which increased the complexity and power consumptionof the communications interface.

A number of signaling methods are known that maintain the desirableproperties of DS, while increasing pin efficiency over DS. One suchmethod is Vector signaling. With vector signaling, a plurality ofsignals on a plurality of wires is considered collectively although eachof the plurality of signals might be independent. Thus, vector signalingcodes can combine the robustness of single circuit DS and the high wirecount data transfer throughput of parallel data transfer. Each of thecollective signals in the transport medium carrying a vector signalingcodeword is referred to as a component, and the number of plurality ofwires is referred to as the “dimension” of the codeword (sometimes alsocalled a “vector”). With binary vector signaling, each component or“symbol” of the vector takes on one of two possible values. Withnon-binary vector signaling, each symbol has a value that is a selectionfrom a set of more than two possible values. The set of values that asymbol of the vector may take on is called the “alphabet” of the vectorsignaling code. A vector signaling code, as described herein, is acollection C of vectors of the same length N, called codewords. Anysuitable subset of a vector signaling code denotes a “subcode” of thatcode. Such a subcode may itself be a vector signaling code. Inoperation, the coordinates of the codewords are bounded, and we chooseto represent them by real numbers between −1 and 1. The ratio betweenthe binary logarithm of the size of C and the length N is called thepin-efficiency of the vector signaling code. A vector signaling code iscalled “balanced” if for all its codewords the sum of the coordinates isalways zero. Additional examples of vector signaling methods aredescribed in Cronie I, Cronie II, Cronie III, Cronie IV, Fox I, Fox II,Fox III, Holden I, Shokrollahi I, Shokrollahi II, and Hormati I.

As previously described, broadband signaling methods partition theavailable information transfer medium in the frequency domain, creatingtwo or more frequency-domain “channels” which may then may transportinformation in a comparable manner to baseband circuits, using knownmethods of carrier modulation to convert the baseband information into afrequency-domain channel signal. As each such channel can beindependently controlled as to amplitude, modulation, and informationencoding, it is possible to adapt the collection of channels to widelyvarying information transfer medium characteristics, includingvariations in signal loss, distortion, and noise over time andfrequency.

Asymmetric Digital Subscriber Line or ADSL is one widely deployedbroadband signaling method used to transport digital data over legacycopper telephony circuits. In ADSL, each of potentially several hundredfrequency-domain channels is independently configured for amplitude,modulation method, and digital carrying capacity, based on theparticular noise and loss characteristics of the copper circuit beingused for transport.

BRIEF DESCRIPTION

Communication of digital information using a combination of baseband andbroadband techniques over multiple wires is described. A four wirecommunications channel having 35 dB of attenuation at 37.5 GHz is usedin provided examples as a typical transport medium for use with thesystems and methods described herein. One embodiment creates twofrequency-based channels over the transport medium, with each channelusing a combination of a vector signaling code and duobinary encoding totransport sets of three data bits over four wires at an effective rateof 56 Gigabits per second per wire.

BRIEF DESCRIPTION OF FIGURES

FIG. 1 illustrates the frequency-domain and time-domain characteristicsof the transport channel model used herein.

FIG. 2 shows the simulated CTLE gain and transmit spectrum of a firstembodiment using ENRZ signaling over a two pair (four wire) transportchannel.

FIG. 3 shows the simulated receive eye opening for the first embodiment.

FIG. 4 shows the simulated CTLE gain and transmit spectrum of a secondembodiment using ENRZ signaling combined with duobinary encoding over atwo pair (four wire) transport channel.

FIG. 5 shows the simulated receive opening for the second embodiment.

FIG. 6 illustrates the spectrums of the broadband and carrier channelsfor the described third embodiment.

FIG. 7 show simulated pulse responses and cross-channel ICI for thethird embodiment.

FIG. 8 shows the simulated receive eyes for the third embodiment.

FIG. 9 is a block diagram of a transmitter embodiment combining basebandand carrier band signaling.

FIG. 10 is a block diagram of an alternative transmitter embodimentcombining baseband and carrier band signaling.

FIG. 11 is a block diagram of a receiver embodiment detecting basebandand carrier band signals.

FIG. 12 shows simulated eye openings for a fourth embodiment, utilizinga baseband plus a single carrier band operating at 224 Gigabits/secondper wire pair.

FIG. 13 shows simulated eye openings for a sixth embodiment, utilizing abaseband plus a single carrier band operating at 112 Gigabits/second perwire pair.

FIG. 14 shows simulated eye openings for a seventh embodiment, utilizinga baseband plus a single carrier band operating at 224 Gigabits/secondper wire pair.

FIG. 15 shows the distribution of data bits and redundancy-augmentedbits across the six subchannels and multiple sequential transmit unitintervals, as described relevant to the ninth and tenth embodiments ofthe invention.

FIG. 16 is a block diagram illustrating the insertion error correctionprocessing into the ++−− carrier subchannel as described for the ninthand tenth embodiments of invention.

DETAILED DESCRIPTION

Interconnection has long been a limiting factor in the design of largedigital systems. Whether at the level of modules interconnected by abackplane, or of functional subsystems interconnected within a largeprinted circuit board, the need for reliable, error free, high-speeddigital interconnection has constantly pushed the limits of availabletechnology to its limits.

The systems and methods described herein provide robust, reliabletransfer of data between at least one transmitting device and at leastone receiving device, at data rates of at least 50 Gigabits per secondper interconnection wire. An example channel model having the frequency-and time-domain characteristics illustrated in FIG. 1 will be used. Itwill be obvious to one familiar with the art that such a transportchannel is incompatible with conventional communication signalingmethods; for example, straightforward NRZ signaling at an example 112Gibabits/second has a Nyquist frequency of 56 GHz, corresponding to anintractable 46 dB attenuation over the proposed physical transportchannel.

This proposed data rate also strains integrated circuit data processingcapabilities within the attached transmitting and receiving devices. Itis therefore presumed that high-speed data handling in these deviceswill be distributed across multiple parallel processing “phases”. As oneexample, rather than a single data path handling data at 100 Gigabitsper second (i.e. with merely 10 picosecond between bits), the same datastream may be distributed across sixteen processing phases, each onethus having a more reasonable 160 picoseconds of processing time perbit. However, this added processing time comes at the cost ofsignificantly increased complexity from the additional processingelements. This distribution of processing also can lead to increasedlatency before a given digital bit result becomes available, limitingthe ability to utilize that result in predicting a subsequent bitresult, which is the basis of the DFE method.

The increasing data transfer rates also lead to physical issues as thewavelength of the propagating signals on the interconnection shrinks. Asone example, the propagating signal wavelength at 56 Gigahertz on aprinted circuit micro stripline is approximately 4 millimeters, thusperiodic anomalies with merely fractional wavelength dimensions (evenincluding the weave of the impregnated fabric comprising the circuitboard) may represent a significant disturbance to signal integrity,stressing available equalization and compensation methods.

Encoding Information Using Hadamard Transforms

As taught in [Cronie I], the Hadamard Transform, also known as theWalsh-Hadamard transform, is a square matrix of entries +1 and −1 soarranged that both all rows and all columns are mutually orthogonal.Hadamard matrices are known for all sizes 2N as well as for selectedother sizes. In particular, the description herein utilizes the 4×4Hadamard matrix as the example encoder.

The order 4 Hadamard matrix used in our examples is:

$\begin{matrix}{H_{4} = \begin{bmatrix}{+ 1} & {+ 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} & {- 1} \\{+ 1} & {+ 1} & {- 1} & {- 1} \\{+ 1} & {- 1} & {- 1} & {+ 1}\end{bmatrix}} & \left( {{Eqn}.\mspace{14mu} 1} \right)\end{matrix}$

and encoding of the three informational bits A, B, C may be obtained bymultiplying those informational bits times the rows 2, 3, and 4 of theHadamard matrix H₄ to obtain four output values, subsequently called“symbol values”. By convention, the results are scaled by an appropriateconstant factor so as to bound the symbol values to the range +1 to −1.It may be noted that the first row of H₄ corresponds to common modesignaling, which is not used herein, with the next three vectors beingused to encode bits A, B, and C respectively into outputs W, X, Y, Z,these vectors also being called “modes” or “subchannels” of the Hadamardcode. As the encoded outputs simultaneously carry information derivedfrom the encoding of A, B, and C, the outputs will be a superposition orsummation of modes, i.e. a sum of the sub-channel code vectors of thevector signaling code.

One familiar with the art will note that all possible values of A, B, Cencoded in this manner result in mode summed values for W, X, Y, Z whichare balanced; that is, summing to the constant value zero. If the modesummed values for W, X, Y, Z are scaled such that their maximum absolutevalue is 1 (that is, the signals are in the range +1 to −1 forconvenience of description) it will be noted that all achievable valuesare permutations of the vector (+1, −⅓, −⅓, −⅓) or of the vector (−1, ⅓,⅓, ⅓). These are called the codewords of the vector signaling code H4.As used herein, this H4 code will subsequently be called Ensemble NRZcode or ENRZ and will be used as a representative example of vectorsignaling code in subsequent examples, without implying limitation.

ENRZ

[Hormati I] teaches that ENRZ has optimum Inter Symbol Interference(ISI) characteristics, and [Holden I] and [Ulrich I] teach it is capableof efficient detection. As previously described, ENRZ encodes threebinary data bits into a four-symbol codeword for transmission, as oneexample, over four wires of a transport medium. If ENRZ signaling isused over four wires of the proposed channel, the data transfer rate maybe achieved with merely a 75 Gigasymbol/second signaling rate,equivalent to 112 Gbps per wire pair, for the two pair transportchannel.

Simulation of a first embodiment combining ENRZ signaling at a 75Gigasymbol/second rate with the reference channel model indicates that atwo tap FFE (transmit Feed-Forward Equalization) may be combined withreceiver continuous-time linear equalization (CTLE) and a 12 tapDecision Feedback Equalizer (DFE), with performance as illustrated inthe graphs of FIG. 2. The receive eye simulation of FIG. 3 shows a 93 mVvertical eye opening and a 14.5 psec edge-to-edge horizontal eyeopening.

Duobinary Encoding

Duobinary encoding is a solution known in the art in which consecutivebits of a serially transmitted data stream are processed to shape andconstrain the resulting transmit data spectrum. It is well known thatInter-Symbol Interference (ISI) such as may be produced by transmissionmedium perturbations will result in the received amplitude of a signalin one unit interval to be perturbed by residual energy from previousunit intervals. As one example, inverted pulse reflections from aperturbation of the transmission medium will cause a received signal tobe reduced by the residual influence of previously transmitted signals.Thus, a transmitter informed of this effect might combine a presentlytransmitted signal value with that of a previous transmission, in anattempt to anticipate or pre-compensate for this inter-symbolinterference effect. Thus, use of partial response codes such asduobinary are often described as a particular form of pre-equalizationfiltering intended to produce constructive ISI, rather than as a literaldata encoding means.

As described in [Beyene], other partial-response codes are known to havecomparable ISI management capabilities. For reference purposes, thecharacteristic equations defining these encodings or filterings arelisted in Table I.

TABLE I Partial Response System Characteristic Equation Duobinaryx_(n) + x_(n−1) Dicode x_(n) − x_(n−1) Modified Duobinary x_(n) −x_(n−2) Class 2 x_(n) + 2x_(n−1) + x_(n−2)

Unless otherwise described, as used herein the duobinary processingperformed is assumed to be a summation of the present and immediatelyprevious transmit unit interval signal, each scaled by a factor of 0.5.Optionally, this may be combined with a transmit lowpass filter tofurther control the transmit spectrum. In other embodiments,ISI-controlling encoding is combined in any order with Hadamardencoding, where the ISI-controlling encoding is any of duobinary,modified duobinary, dicode, class2, or a Hamming filter as subsequentlydescribed. In such embodiments, the ISI-controlling encoding may also bedescribed as being performed by a partial response encoder, embodyingany of the partial response encodings or filterings above.

If the characteristics of the communications channel are extremely wellunderstood, it may be possible to configure the ISI-controllingoperation of the transmitter such that no explicit complementaryoperation is required at the receiver, the effective action of thechannel characteristics themselves serving to perform the inverseoperation. Other embodiments may explicitly detect, as one example, theternary signals produced by duobinary encoding of binary data, followedby an explicit duobinary to binary decoding operation. Alternatively,commonly used receiver ISI elimination techniques such as DFE will alsoefficiently address the effects of such transmitter ISI compensation. Aseach example receiver in this document already incorporates DFE, nofurther receiver duobinary (or other partial response code) processingwill be shown.

A second embodiment incorporating ENRZ encoding at a 75Gigasymbol/second rate, subsequent duobinary processing of each wiresignal, a 2 tap FFE, CTLE, and a 12 tap DFE was simulated using thereference channel model, producing the CTLE gain and spectrum resultsshown in FIG. 4. The receive eye simulation shown in FIG. 5 shows a 75mV vertical receive eye opening and a 13.7 psec edge-to-edge horizontaleye opening.

These results, although representing considerable improvement overstraightforward NRZ data transmission, indicate additional work isneeded.

Channelization

If purely baseband communications solutions are insufficient, might abroadband approach be of benefit? Historically, such significant levelsof physical transport channel limitation had been seen and addressedbefore, albeit at far lower data rates, during the efforts to providehigh speed digital services over the legacy copper wire infrastructureof the telephony network. For DSL at its desired 3 Megabit data rate, apropagating signal wavelength was several hundred meters, whichcorrelated strongly with the typical spacing of wire stubs, splices, andinsulation abrasions seen in the field. Thus, an uncompensated frequencyresponse for a typical copper telephony signal path would exhibitnumerous notches and slopes caused by reflective interference amongthose anomalies, dissipative attenuation from degraded wires andinsulation, and intrusive noise from sources such as AM radiotransmitters.

Ultimately, multichannel frequency domain channelization was used toconstrain the effect of those legacy transport issues. One commonlydeployed Asymmetric Digital Subscriber Line (ADSL) solution, forexample, partitioned the approximate 1 MHz of available transport mediumbandwidth into 4.3125 kHz channels. Each channel was then independentlytested for attenuation and signal-to-noise ratio, with different datathroughput rates assigned to each channel depending on those testresults. Thus, a channel frequency coinciding with a frequency responsenotch or significant external noise source would not be used, whileother channels not presenting those issues could be used at fullcapacity. Unfortunately, the generation and detection of such a highchannel count protocol relies on the availability of inexpensive digitalsignal processing solutions, and such technology has scaled inperformance over time by perhaps a factor of ten, versus the approximatefactor of 100,000 data rate increase in the present application.

Thus, although the present channel attenuation issues suggest abroadband approach may be useful, the conventional high-channel-countembodiment methods known to the art are incompatible with theanticipated data rate. A new approach specifically designed for highspeed processing will be required.

Broadband Duo Binary ENRZ

A third embodiment combines ENRZ, duobinary, and a two frequency-domainchannel approach to address the issues of the previous proposals. Thefirst frequency channel is at baseband, i.e. comparable to the singlechannel of the previous embodiment. The second frequency channel iscomposed of the same ENRZ+duobinary signaling modulating a sinusoidalcarrier, chosen to minimize the frequency overlap between spectralcomponents of the baseband and of the carrier channel.

In the following example, a carrier frequency of 37.5 GHz will be used,with no limitation implied. Comparable results have been obtained insimulations using a 30 GHz carrier frequency, and lower frequencies maybe used with improved channel attenuation characteristics but somewhathigher inter-channel interference, as will be shown in a subsequentexample.

Both frequency channels run at a signaling rate of 37.5 Gsymbols/second,with three data bits being transported over the four wires of thebaseband channel, and a second three data bits being transported overthe same four wires using the carrier channel, to produce an aggregatethroughput equal to the previous embodiments. With the same datathroughput distributed across two channels, the required signaling rateper channel is halved, thus potentially allowing a much wider horizontaleye opening.

FIG. 6 illustrates the spectrums of the broadband and carrier channelsand the corresponding pulse shapes of the two channel signals, asproduced by a simulation of this embodiment operating over the referencechannel model.

In this embodiment, data for each of the two channels is separately ENRZencoded, and then each of the four signaling streams carrying the ENRZcodewords is duobinary encoded by summing the present and immediatelyprevious Unit Interval's value, each scaled by a factor of 0.5.(Alternatively, the summation of the values may subsequently be scaledby the same factor, or the scaling may be subsumed into lateramplification and/or filtering functions.) Each of the two resultingduobinary encoded streams, herein also referred to as sets ofbaseband-encoded symbols, are pre-emphasized using a two tap FFE, thenpassed through a Butterworth lowpass filter of order 2 with a cutofffrequency of 9.37 Gigahertz for spectral shaping and ICI reduction. Thefiltered stream for the carrier channel modulates a sinusoidal carrierat 37.5 GHz, the result of which is linearly combined with the filteredstream for the baseband channel for transmission over the transportchannel.

As the subchannels of a Hadamard code such as ENRZ are linear, that is,they transparently communicate non-binary as well as binary signals, theorder in which duobinary and ENRZ encoding is performed may be reversed.In at least one such alternative embodiment, each of the three data bitsis separately duobinary encoded before being presented to the ENRZencoder, rather than the ENRZ code outputs being duobinary encoded, foreach of the baseband and carrier channels.

Transmitter

FIG. 9 is a block diagram of one embodiment of a Broadband DuobinaryENRZ transmitter. Data at an aggregate rate of 224 Gigabits/secondenters MUX 910, which separates it into two independent data streams 915and 918, each of 112 Gigabits/second that serve as data inputs to thebaseband and carrier channels.

The baseband channel data is ENRZ encoded 920, with each three bits ofinput data producing one code word of four symbol values. Each basebandsymbol value will subsequently be processed independently and ultimatelytransported (along with its comparable carrier channel processed symbolvalue) on its own wire. Processing for each baseband symbol value mayinclude duobinary encoding by partial-response signaling encoder 940 andlow-pass filtering and amplification by amplifier 960 as needed to meetsystem signal level criteria, to produce a processed baseband output. Insome embodiments, the partial response signaling encoder may beimplemented with two sets of analog voltage generators, where each setis alternately driven with a codeword input and provides a set ofvoltages representing the codeword symbols, but the generators maintaintheir outputs for a duration of two signaling intervals. The sets ofvoltages are summed at a signal summing circuit. While each set ofvoltages changes at ½the symbol rate, because they are staggered intime, the outputs of the summing circuit change at the symbol rate, andrepresent the sum of the current symbol and the prior symbol. In someembodiments, the encoder such as ENRZ encoder 920 may comprise twoencoders also operating at ½ rate, each encoder configured to drive acorresponding set of analog voltage generators.

Processing for the carrier channel is comparable to that of the basebandchannel to the point of carrier modulation, with carrier channel data918 being ENRZ encoded 930, with each three bits of input data producingone codeword of four symbol values. Each carrier symbol value willsubsequently be processed independently, and then mixed with itscomparable processed baseband symbol value for wire transmission.Processing for each carrier symbol value consists of duobinary encoding950, low-pass filtering and amplification 970 as needed to meet systemsignal level criteria, and modulation 980 of the 37.5 GHz Carrier toproduce a processed and modulated carrier output.

Each of the four processed baseband outputs is summed 990 with itscomparable processed and modulated carrier outputs, producing wireoutputs identified in FIG. 9 as Wire A, Wire B, Wire C, and Wire D.

FIG. 10 shows an alternative transmitter embodiment, in which duobinaryencoding 1020 and 1030 is performed prior to ENRZ encoding 1040 and1050. Other than the order of these operations, this alternativetransmitter is identical to that of the embodiment of FIG. 9.

Receiver

One embodiment of a comparable Broadband Duobinary ENRZ receiver isshown in the block diagram of FIG. 11. Each wire signal from thetransport medium Wire A, Wire B, Wire C, and Wire D is amplified andfrequency equalized by a continuous-time linear equalizer (CTLE) 1110,and then the four amplified and equalized received signals are input tothree linear ENRZ mixers 1120. In some embodiments, CTLEs 1110 mayinclude analog delay circuits, and the receiver may include a skewcontrol circuit 1112 configured to provide a skew control signal to eachof the CTLEs 1110. In some embodiments, the analog delay circuits may beall-pass filters (including a switched capacitor bank, for example)configured to adjust an analog delay of each individual wire A-D. Insome embodiments, the skew control circuit 1112 may be configured tooperate on the outputs of samplers 1180 that operate on the passband MICoutputs in order to determine a skew control signal for adjusting analogdelay values of each wire, however this should not be consideredlimiting. In one embodiment, each sub-channel MIC may be evaluated byadjusting decision thresholds, and responsively measuring an effectiveeye opening, and then individual wire skews may be adjusted in order toincrease the effective eye opening. In some embodiments, the sub-channelMIC with the narrowest effective eye opening is adjusted first. Further,alternative analog delay circuits known to those of skill in the art maybe implemented.

As taught by [Holden I], such ENRZ receive mixing is commonly utilizedat baseband by so-called multi-input comparators (MIC) to detect ENRZcodewords. Here, the ENRZ mixing in such MICs produces three linearsignal “subchannels” comprising a linear superposition of baseband andbroadband, or carrier-modulated, results for each of the two ENRZencoded streams. The mixing operations are defined as:

R ₀=(A+C)−(B+D)  (Eqn. 2)

R ₁=(C+D)−(A+B)  (Eqn. 3)

R ₂=(A+D)−(B+C)  (Eqn. 4)

where R₀, R₁, R₂ are the three resulting linear signal channels outputfrom ENRZ mixers 1120, and A, B, C, D are the four received wire signalsoutput from the CTLE 1110. Equivalent mixing results may be obtainedusing other algebraic permutations of these equations as may be producedby a different ordering of wire labels; as one example R₁=(A+B)−(C+D) isequivalent to Eqn. 3 if the wires are labeled in reverse order. MICsembodying such mixing results may also be identified by the signs ofwire terms in their defining equation, e.g. ++−− for this example.

A four pole Butterworth lowpass filter 1130 with a cutoff frequency of18.75 GHz is used to extract the baseband component from each of thelinear signal subchannels. As is common practice in the art, the signalamplitude of each of the linear signal subchannels is measured orcaptured at a particular moment or interval of time by samplers 1140 at37.5 Giga sample/second rate to produce the three decoded Baseband Dataout bits, at an aggregate 112 Gigabit/second data rate. Concurrently,each decoded bit is presented to a DFE computation 1150, producing a DFEcorrection signal used to adjust that bit's sampler threshold. DigitalFeedback Equalization is well known in the art, thus will not be furtherdescribed here, other than noting that each DFE computation 1150 isindependent, and will provide both correction oftransport-channel-induced ISI and of intentionally generated transmitterISI compensation.

It should be noted that the described DFE correction operating onsubchannels of the vector signaling code is distinct from the commonart, where which DFE correction is performed on e.g. received wiresignals. As the history maintained by the DFE must accurately representthe values of each unit interval in the history, a conventional DFEwould have to maintain ternary, quaternary, or higher-order historyvalues to represent a vector signaling code having 3, 4, or morepossible symbol values. In contrast, binary data communicated over avector signaling code subchannel requires maintenance of merely a binaryhistory using the described DFE correction.

Simultaneously, a second order Butterworth high pass filter 1150 with acutoff of 37.5 GHz extracts the carrier channel information from thethree linear signal subchannels. Balanced mixers 1160 provided with a37.5 GHz carrier signal converts these modulated signals back tobaseband where, as with the baseband channel signals, a four poleButterworth lowpass filter 1070 with a cutoff frequency of 18.75 GHz isused followed by sampling 1080 at 37.5 Gig sample/second rate on each ofthe subchannels to produce the three decoded Carrier Data out bits, atan aggregate 112 Gigabit/second data rate. As with the baseband data,each decoded carrier data out bit is presented to a DFE computation1190, producing a DFE correction signal used to adjust that bit'ssampler threshold. Each DFE computation 1190 is independent, and willprovide both correction of transport-channel-induced ISI and ofintentionally generated transmitter ISI compensation.

Because of the significant frequency-dependent loss characteristics ofthe transport channel, the gain of the receive baseband channel is setto 14 dB, while the gain of the carrier channel is set to 26 dB.Similarly, the transmitter gain for the carrier channel is set to 3times that of the baseband channel to provide pre-emphasis.

Simulated pulse responses and cross-channel ICI for this embodiment areshown in FIG. 7, assuming two taps of transmit FFE and fifteen taps ofreceive DFE. Receive eyes for the baseband and carrier (passband)channels are shown in FIG. 8. Eye openings are 54 mV vertical and 24.1psec horizontal for the baseband, and 56 mV vertical and 38.7 psechorizontal for the passband, a considerable improvement over theprevious embodiments.

Skew Considerations

As with any vector signaling code solution, skew must be constrainedacross the transport paths carrying symbols of the same codeword, as thecodeword must be presented as a coherent whole to the receiver'sdetector to be properly recognized. Roughly speaking, propagationlatencies across the various transport paths must be matched to lessthan one half the expected eye width to permit detection, and betterthan that value to avoid eye width degradation. Known approachesincluding introduction of variable delay lines and/or FIFO buffers forpath compensation, separate CDR and sample timing for individual wires,and transmit-side pre-skew compensation. However, these techniques mustbe applied cautiously, as they may also lead to increased inter-symbolinterference, transmit simultaneous switching noise, and higherperceived receive common mode signals.

Because the baseband and carrier-band channels carry separate ENRZencoded data and are separately receive sampled, their data streams maybe considered to be independent and thus do not require absolutetemporal alignment. This is an advantage, as differences between thefiltering characteristics of the two channels will introduce differenttime delays, which inherently introduces a timing difference between theset of data bits received at baseband, and the set of data bits receivedat carrier band. As will be apparent to one familiar with the art, thesesets of bits may be passed through retiming latches, FIFO buffers, orother known means to align them with a common timing reference.

Alternative Embodiments

A number of variations to the preceding embodiments have beenconsidered, all within the scope of the described invention. Transmitsignal generation of the ENRZ symbol values, their ISI-controllingencodings, or both may be produced using Digital to Analog convertershaving an appropriate number of bits. Similarly, mixing of broadband andcarrier signals within the transmitter may be done digitally.

Transmitter and receiver embodiments may incorporate additional gainand/or frequency-dependent filtering stages to meet the describedvertical eye openings, or to compensate for channel characteristicsdiffering from those of the reference channel model. Particularamplitudes, gains, attenuation characteristics, etc. are provided fordescriptive purposes, without implying limitation.

At least one embodiment performs additional prefiltering of signalswithin the transmitter to zero out the first few pre-cursors of thechannel, thus avoiding the need for extensive DFE tap unrolling at thereceiver.

The example broadband receiver embodiment described converts thecarrier-based channel to baseband for subsequent detection. Thispresumes that the local carrier available at the receiver is coherentwith the transmitter's carrier signal, and is thus derived using aPhase-locked loop or other known method. Other known art receivermethods are well known and may also be incorporated in alternative andequivalent embodiments.

A receiver embodiment may also utilize Analog-to-Digital samplingfollowed by some or all of the previously-described filtering, mixing,and sampling being performed using digital signal processing methods.

Extension to Higher Data Rates

The embodiments described herein may be extended to support data ratesof 224 Gigabits per second per wire pair.

In a fourth embodiment incorporating such extension, the data isprefiltered at the transmitter to add more controlled ISI. As oneexample, a Hamming filter of order 7 is used having the coefficients:

H=[0.02,0.09,0.23,0.30,0.23,0.09,0.02]  [Eqn. 5]

This is contrasted with the duobinary encoding of the previous examples,which corresponds to a transmit filter with the coefficients:

H=[0.5,0.5]  [Eqn. 6]

In this fourth embodiment the data rate in each of the baseband andcarrier channels is doubled, to 75 Gigasymbols/second, resulting in anaggregate data throughput equivalent to 112 Gigabits per second perwire, or 448 Gigabits per second for the four wire interconnection.Simulated eye openings are shown in FIG. 12, where the baseband channelhas 93 mV of vertical and 8.3 psec of horizontal eye opening, and thecarrier channel has 42 mV of vertical and 16.6 psec of horizontal eyeopening, assuming 3 pre-cursor taps of transmit equalization, and 15taps of receive DFE.

Alternatively, an embodiment may utilize additional carrier channels. Asone example, a baseband channel plus three carrier channels operating atcarrier frequencies chosen to minimize the frequency overlap betweenspectral components of the various channels may be combined, with eachchannel carrying a data stream combining ENRZ encoding with anISI-controlling encoding with each channel operating at a rate of 37.5Gigasymbols/second as previously described.

Extension to Other Base Signaling Schemes

As previously noted, the embodiments described herein may be used withunderlying vector signaling codes other than ENRZ, which has been usedfor purposes of description in the previous examples without implying alimitation. Other multi-wire signaling schemes may also be combined withthe described ISI management and channelization techniques, as should beunderstood by anyone of ordinary skill in the art.

For example, a fifth embodiment is identical to that of the previouslydescribed fourth embodiment, except that differential signaling is usedon each two wire pair at a signaling rate of 75 Gigabits/second/pair,rather than ENRZ across all four wires. Data on each channel isprefiltered at the transmitter to add more controlled IS using a Hammingfilter of order 7 having the coefficients:

H=[0.02,0.09,0.23,0.30,0.23,0.09,0.02]  [Eqn. 7]

In this fifth embodiment the aggregate throughput is thus 300Gigabits/second; 75 Gigabits/second per wire pair for two wire pairs,for each of the two channels.

Use of a Lower Carrier Frequency

As previously mentioned, a lower carrier frequency may be used to bringthe carrier-modulated channel into a lower attenuation region of thetransport channel model, at the cost of increased inter-channelinterference.

A sixth embodiment operates with a baseband channel and one carrierchannel modulating a carrier frequency of 19.5 GHz. Both baseband andcarrier channels utilize ENRZ encoding and Duobinary filtering, aspreviously described, at a signaling rate of 37.5 GBaud, equivalent to a26.66 psec UI. The resulting signal spectrum experiences a 15 dB channelloss at baseband, and a 30 dB loss at the carrier channel. Thesimulation results shown in FIG. 13 and summarized in Table 2 are basedon 600 mV Tx amplitude, 200 uV RMS channel noise, a 1:7 baseband tocarrier channel power ratio, 1 pre- and 1 post-cursor TX FIR, up to 12dB of Rx CTLE, and 12 taps of Rx DFE. Eye openings sufficient to obtainat least a 10E-6 Bit Error Rate (BER) were observed.

TABLE 2 Band MIC Vertical mV Horizontal psec % UI Carrier ++−− 3.9716.66 62.5 Channel +−+− 5.87 20.21 75.8 +−−+ 5.87 20.21 75.8 Baseband++−− 6.64 17.29 64.9 +−+− 6.43 17.08 64.1 +−−+ 6.45 17.08 64.1

For descriptive convenience, the three ENRZ subchannels on each of theCarrier and Baseband frequencies are identified by the logical wirecombinations comprising the defining equation of their correspondingmulti-input mixer. Thus, as one example, the mixed combination of wiresA, B, C, D corresponding to the mixer performing the (A+B)−(C+D)operation is identified as ++−− in Table 2.

As may be seen in FIG. 13 and Table 2, the eye opening for the ++−−carrier subchannel is significantly smaller than the other eyes, and isthus the limiting factor on performance. In particular, the reducedhorizontal eye opening indicates that subchannel may be significantlyimpacted by wire skew in the transport channel.

Incorporation of Error Correcting Codes

A seventh embodiment operates with a baseband channel and one carrierchannel modulating a carrier frequency of 18.5 GHz. Both baseband andcarrier channels utilize ENRZ encoding and order 11 Hamming filtering,at a signaling rate of 75 GBaud, equivalent to a 13.33 psec UT. Theresulting signal spectrum experiences a 14 dB channel loss at baseband,and a 22 dB loss at the carrier channel. The simulation results shown inFIG. 14 and summarized in Table 3 are based on 800 mV Tx amplitude, 200uV RMS channel noise, 260 femto-seconds of random jitter (Rj), a 1:7baseband to carrier channel power ratio, 1 pre- and 1 post-cursor TXFIR, up to 12 dB of Rx CTLE, and 25 taps of Rx DFE.

TABLE 3 Band MIC Vertical mV Horizontal psec % UI Carrier ++−− 1.76 8.6564.9 Channel +−+− 3.02 10.52 78.9 +−−+ 2.93 10.31 77.3 Baseband ++−−2.86 9.48 71.1 +−+− 2.74 9.38 70.4 +−−+ 2.72 9.38 70.4

As with the previous example, eye openings sufficient to obtain a 1E-6BER were observed, with the ++−− carrier subchannel again limiting theoverall performance, especially in the presence of transport channelwire skew.

Various approaches were considered to mitigate this subchannel limitingperformance, allowing improved system BER to be achieved.

An eighth embodiment is identical to the previously described seventhembodiment, but the marginal ++−− carrier subchannel is not used totransmit data. This results in an overall throughput of 5*75=375 Gbpsover the four wire transport medium, equivalent to an effective 187.25Gbps per wire pair.

A ninth embodiment is identical to the previously described seventhembodiment, with an additional reliability protocol imposed on datatransmitted over the marginal ++−− carrier subchannel. As one exampleoffered without limitation, a “send three times” reliability protocolmay be used on that subchannel to transmit the same data bit in threeconsecutive UIs, with a majority detector used at the receiver toidentify the received data bit. Thus, this embodiment transmits a totalof 16 bits (rather than the seventh embodiment's 18) in three Ms. Thisresults in an overall throughput of 6*75*(16/18)=400 Gbps over the fourwire transport medium, equivalent to an effective 200 Gbps per wirepair. Addition of this reliability protocol provides an effective BER of1E-6 if the underlying subchannel provides at least a 5.7E-4 BER,equivalent to an improvement of the vertical eye by 6 dB and almost adoubling of the horizontal eye opening.

A tenth embodiment is identical to the previously described seventhembodiment, with a Forward Error Correcting protocol imposed on datatransmitted over the marginal ++−− carrier subchannel. As one exampleoffered without limitation, four consecutive data bits may be encodedusing a [7,4,3] Hamming code to produce seven Hamming encoded bits to besequentially transmitted over that subchannel in seven UIs, with thecorresponding Hamming decoder used at the receiver to recover thereceived data bits. Thus, this embodiment transmits a total of 39(rather than the seventh embodiment's 42) data bits in seven consecutiveUIs, resulting in an overall throughput of 6*75*(39/42)=417.86 Gbps.equivalent to an effective 208.93 Gbps per wire pair. Addition of thisFEC encoding provides an effective BER of 1E-6 if the underlyingsubchannel provides at least a 3.6E-3 BER, equivalent to an improvementof the vertical eye opening by 7 dB and an 2.5×enlargement of thehorizontal eye opening.

This distribution of data bits and redundancy-augmented bits across thesix subchannels and multiple sequential transmit unit intervals asdescribed relevant to the ninth and tenth embodiments of the inventionis illustrated in FIG. 15.

FIG. 16 is a block diagram showing error correction being added to anencoded transmission subchannel and the corrected data identified at thereceiver. At the transmitter, Data In is distributed 910 among thecarrier subchannels and the baseband subchannels, as previously shownrelative to FIG. 9 and FIG. 10. The portion of the data bits directed tothe ++−− carrier subchannel are passed through an error correctionfunction 1510 which increases its redundancy; relative to the ninthembodiment this redundancy is obtained via repetition, relative to thetenth embodiment this redundancy is obtained via a Hamming Code encoder.The data bits directed to the carrier subchannels 915 and the data bitsdirected to the baseband subchannels 918 are then processed aspreviously described in FIG. 9 or FIG. 10. At the receiver, data fromthe sampler associated with the ++−− mixer carrier channel is directedto error correction function 1520, which identifies the original databits; a majority detector is used relative to the ninth embodiment, anda Hamming Code decoder is used relative to the tenth embodiment. Theoriginal data bits from 1520 and the sampler outputs from the othersubchannels may be combined 1530 to produce an aggregated received datastream identical to that presented to the transmitter.

It will be obvious to one skilled in the art that redundancy and/orforward error correction may be applied to more than one subchannel,with a corresponding improvement in that subchannel's effective eyeopening but also resulting in decreased delivered data rate due to theinevitable overhead. Thus, these examples applying such solution to asingle subchannel should not be considered as limiting, but may bepreferred within the parameters of the example.

1. A method comprising: receiving, at a plurality of orthogonalsubchannel multi-input comparators (MICs), a set of wire-specificinputs, each wire-specific input carrying a combination of a respectivebaseband symbol of a baseband codeword and at least one respectivecarrier-modulated symbol of at least one carrier-modulated codeword; andgenerating a plurality of superposition subchannel signals, eachsuperposition subchannel signal generated by a corresponding orthogonalsubchannel MIC forming a respective subchannel-specific linearcombination of the set of wire-specific inputs, each superpositionsubchannel signal comprising a respective superposition of (i) abaseband subchannel signal and (ii) one or more carrier-modulatedsubchannel signals, the baseband subchannel signal and the one or morecarrier-modulated subchannel signals associated with the basebandcodeword and the at least one respective carrier-modulated codeword,respectively.
 2. The method of claim 1, further comprising generating aset of decoded output bits based on the plurality of superpositionsubchannel signals.
 3. The method of claim 2, wherein generating the setof decoded output bits are generated using a plurality of samplers. 4.The method of claim 3, further comprising applying decision feedbackequalization (DFE) correction signals to adjust thresholds of theplurality of samplers.
 5. The method of claim 1, further comprisingparsing, using a plurality of filters, each superposition subchannelsignal of the plurality of superposition subchannel signals into thebaseband subchannel signal and the at least one carrier-modulatedsubchannel signal.
 6. The method of claim 1, further comprisinggenerating, for each of the at least one carrier-modulated subchannelsignal, a corresponding demodulated-baseband subchannel signal using abalanced mixer circuit of a set of at least one balanced mixer circuit.7. The method of claim 1, wherein each respective subchannel-specificlinear combination is performed according to a row of an orthogonalmatrix.
 8. The method of claim 7, wherein the orthogonal matrix is aHadamard matrix.
 9. The method of claim 8, wherein the Hadamard matrixhas a size of
 4. 10. The method of claim 1, wherein the basebandcodeword and the at least one carrier-modulated codeword are balanced.11. An apparatus comprising: a plurality of wires of a multi-wire bus,the plurality of wires configured to carry a set of wire-specificinputs, each wire-specific input comprising a combination of arespective baseband symbol of a baseband codeword and at least onerespective carrier-modulated symbol of at least one carrier-modulatedcodeword; and a plurality of orthogonal subchannel multi-inputcomparators (MICs) configured to generate a plurality of superpositionsubchannel signals, each superposition subchannel signal generated by acorresponding orthogonal subchannel MIC forming a respectivesubchannel-specific linear combination of the set of wire-specificinputs, each superposition subchannel signal comprising a respectivesuperposition of (i) a baseband subchannel signal and (ii) one or morecarrier-modulated subchannel signals, the baseband subchannel signal andthe one or more carrier-modulated subchannel signals associated with thebaseband codeword and the at least one respective carrier-modulatedcodeword, respectively.
 12. The apparatus of claim 11, furthercomprising a plurality of samplers configured to generate a set ofdecoded output bits based on the plurality of superposition subchannelsignals.
 13. The apparatus of claim 12, further comprising a decisionfeedback equalization (DFE) circuit configured to generate a set of DFEcorrection signals based on the set of decoded output bits.
 14. Theapparatus of claim 13, wherein the DFE correction signals adjustsampling thresholds of the plurality of samplers.
 15. The apparatus ofclaim 11, further comprising a plurality of filters configured to parseeach superposition subchannel signal of the plurality of superpositionsubchannel signals into the baseband subchannel signal and the at leastone carrier-modulated subchannel signal.
 16. The apparatus of claim 11,further comprising a set of at least one balanced mixer circuits, eachbalanced mixer circuit configured to operate on a respectivecarrier-modulated subchannel signal and to responsively generate acorresponding demodulated-baseband subchannel signal.
 17. The apparatusof claim 11, wherein each respective subchannel-specific linearcombination is performed according to a row of an orthogonal matrix. 18.The apparatus of claim 17, wherein the orthogonal matrix is a Hadamardmatrix.
 19. The apparatus of claim 18, wherein the Hadamard matrix has asize of
 4. 20. The apparatus of claim 11, wherein the baseband codewordand the at least one carrier-modulated codeword are balanced.